Energy Recovery From The Leakage Inductance Of The Transformer

ABSTRACT

Electronic circuitry and method of operating the same to shape and reduce the circulating current through the active clamp in a flyback converter and to harvest most of the leakage inductance energy to provide the bias power. Methodologies for minimizing the circulating energy in the clamp circuit in order to improve efficiency of operation of the same. A method for using a portion of the leakage inductance energy in order to create zero voltage switching conditions at the main primary switch.

CROSS-REFERENCE TO RELATED APPLICATIONS

This patent application claims priority from and benefit of the U.S.Provisional Patent Application Ser. No. 62/571,594 filed on Oct. 12,2017, the disclosure of which is incorporated herein by reference.

TECHNICAL FIELD

This invention pertains to electronic devices employing a powerconverter configured around a flyback topology and, in particular, topower converters and related methodologies directed to improvements ofoperation of such power converters achieved as a result of gainfulharvesting and various usage, of energy of the leakage inductance of theflyback topology, in related electronic circuits.

BACKGROUND

The flyback topology is, arguably, one of the most used circuittopologies in the field of power conversion, especially in lower- tomedium-power applications (such as AC-DC adapters, for example). Thereason for high level of utilization of the flyback topology is rootedin its simplicity and low cost of implementation, as well as in the factthat the so-configured electrical circuitry can operate efficiently overa very large range of input voltage. In AC-DC adapter applications withpowers under about 70W, in order to gain a capability to be applicablesubstantially universally all over the worlds, the circuits formattedaccording to the flyback topology are used to operate after an outputfrom a simple bridge rectifier, while the alternating-current inputvoltage ranges from 90 Vac to 264 Vac. (Conventionally, a rectifier isknown an electrical device that converts alternating current to directcurrent, which flows in only one direction.)

To meet all the AC-voltage standards for different countries, whenplaced at the output of the rectifier, in the flyback converter has tobe able to operate efficiently with a DC input voltage ranging from 127Vdc to 375 Vdc (which is a range in which the ratio of the upper inputvoltage limit to the lower voltage limit is almost 3:1.) In addition tothat, the new standards for power delivery require that the adaptersprovide a voltage output ranging from 5V to 20V(with the ration of theupper voltage limit to the lower voltage limit of 4:1, as far as theoutput voltage is concerned). Most of the forward-derived topologies(such as, for example, half-bridge topology, two-transistor forwardtopology, full bridge topology, to name just a few) are not able tooperate efficiently over such large input and output voltage rangesprovided by the transfer function of the flyback topology based circuit.

The trend for miniaturization of portable equipment (for example,portable computing devices such as laptops and tablets, for example)extends this demand even further, as a result of which the AC-DCadapters also became subject to these requirements. Presently, most ofthe laptops and tablets require, for operation, power ranging from 30 Wto 65 W. The significant technological advancement in portable computingdevices, the size of laptops and tablets has been significantly reducedwhile the AC-DC adapters used to power such devices remain quite large(for example, dimensions of a typical adaptor for a small tablet deviceare about 3.3″ by 1.8″ by 1.3″ or so). This has created pressures forthe size reduction of the AC-DC adapters. An ability to reduce the sizeof the required adapters while maintaining the convection-based coolingmethodology used today requires some significant improvement inefficiency of the adapters as well as decrease of size of the magneticand capacitive storage elements.

Over the years, the efficiency of the AC-DC adapters has been increasedfrom about 70% to about 89-90% (in the most recent products such as theApple 30 W adapter, for example), mostly due to the significant progressin semiconductor industry and a better understanding of magnetictechnology. The flyback topology, however, possesses several drawbacksthat limit its efficiency of operation. In most of the application theflyback-topology circuitry operates in a discontinuous mode. In adiscontinuous mode of operation, the magnetizing current is first builtup from zero to a peak level during the time period when the main switchis conducting; and after the main switch turns off, the magnetizingcurrent flows into the secondary side winding and transfers the energyto the output capacitor until the value of the magnetizing currentdecreases to zero. This portion of the operation cycle is followed by asecond period of time, referred as “dead time”, when no energy is storedin the transformer or transferred to the secondary. Together, the firstand second period of time characterize the discontinuous mode ofoperation of the flyback-topology circuits. When the “dead time” isreduced to the transition time which is the time interval wherein thevoltage across the main switch decays from the level it had during thetime when the magnetizing current flows into the secondary winding toits lowest level which occurs in the beginning of “dead time”, this modeof operation is referred as a critical conduction mode of operation.

This disclosure presents several electronic-circuitry configurationsthat address the limitations conventionally associated with the flybacktopology. The proposed solutions increase the efficiency of theflyback-topology-utilizing power converters above about 94%, decreasethe level of dissipated heat and, as a result, produce a much higherpower density (for example, above 27 W/in³).

SUMMARY

Embodiments of the invention provide an electronic circuitry havingprimary and secondary sides and including (i) a flyback power converterthat has an input voltage source; a transformer having primary andsecondary windings, on the primary and secondary sides, respectively; amain switch in series with the primary winding on the primary side, anda synchronous rectifier in series with the secondary winding on thesecondary side, and (ii) an active clamp circuit across the main switch,the active clamp circuit containing a clamp switch and the clampcapacitor in series with the clamp switch. Here, the clamp switch isconfigured to be turned on at a moment of time after the main switch isturned off, and to be turned off at a moment of time prior to the momentof time at which current passing through the secondary winding reaches azero level. The electronic circuitry is characterized by a first valueof rms current through the clamp capacitor. In a related embodiment, theelectronic circuitry further incudes an auxiliary circuit that containstwo additional rectifiers connected in parallel with one another and inseries with an electronic component configured to store electromagneticenergy (where a cathode of a first of the two additional rectifiers isdirectly electrically connected with a cathode of the active clampcircuit; where an anode of a second of the two additional rectifiers isdirectly electrically connected with the cathode of the first of the twoadditional rectifiers; where a cathode of the second of the twoadditional rectifiers is directly electrically connected with a firstterminal of said electronic component; and where a second terminal ofthe electronic component is electrically connected with an anode of thesecond of the two additional rectifiers). Such auxiliary circuit, whenadded, is configured to reduce the rms current through the clampcapacitor from the first value to a second value, the second value beingat least 40% lower than the first value.

A method for operating the above-identified electronic circuitryincludes electrically-connecting the circuitry with an auxiliaryelectronic circuit in series with the clamp capacitor, where theauxiliary circuit contains two rectifiers and an auxiliary energystorage; and directing a current, flowing through a leakage inductanceof the primary side in operation of said circuitry, to flow through theclamp capacitor and then through a first of the two rectifiers towardsthe auxiliary energy storage to change said first charge value to asecond charge value. Here, the second charge value is smaller than thefirst charge value and a current passing through the clamp capacitorafter the mains switch is turned off is a clamp capacitor current.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 depicts a conventional flyback topology utilizing synchronousrectification.

FIG. 2 illustrates the key waveforms of operation of a standard flybackconverter.

FIG. 3 schematically illustrates a flyback converter with an activeclamp.

FIG. 4 shows plots representing the key waveforms of operation of aflyback converter with an active clamp.

FIG. 5 shows the key waveforms of operation of a flyback converter withan active clamp.

FIG. 6 shows the key waveforms of operation of the flyback topology withan active clamp, where the clamp circuit is controlled to increase theenergy injection into the resonant circuit formed by the primaryinductance of the transformer and the parasitic capacitance reflectedacross the primary switch.

FIG. 7 illustrates the key waveforms of operation of an embodimentincluding the flyback topology with an active clamp where the clampswitch is turned off prior to the moment when the current passingthrough the secondary winding reaches zero, and where the currentpassing through the clamp circuit is zero when the clamp is turned off

FIG. 8 presents the key waveforms of operation of the flyback topologywith an active clamp where the clamp switch is turned on for apredetermined period of time prior the moment when the current in thesecondary winding reaches zero.

FIG. 9 illustrates the key waveforms of operation of an embodimentincluding the flyback topology with an active clamp, where the clampswitch is turned on for a predetermined period of time prior to themoment when the main switch is turned on.

FIG. 10 schematically illustrates an embodiment including a flybacktopology with an active clamp, configured to harvest the energy of theleakage inductance (for example, in order to power the bias of theconverter and to have extra energy transferred to the secondary side, asdiscussed below). Here, an additional circuit formed by two rectifiers(shown as diodes D1, D2 in this example) and a voltage source Vb isplaced in series with the clamp circuit.

FIG. 11 illustrates the key waveforms of an embodiment including theflyback topology with an active clamp, incorporating the current-shapingand the energy-recovery circuit of FIG. 10.

FIG. 12 depicts the key waveforms of the flyback topology with an activeclamp incorporating the current-shaping and the energy-recovery circuitof FIG. 10 as applied to the circuit the operation of which is depictedin FIG. 9.

FIG. 13 depicts a flyback topology with an active clamp from FIG. 10,where a portion of the energy from the leakage inductance is transferredto the bias circuit.

FIG. 14 illustrates the circuitry utilizing a flyback topology with anactive clamp (from FIG. 10), in which the clamp switch is self-drivenfrom the main transformer.

FIG. 15 illustrates the circuitry utilizing a flyback topology with anactive clamp (from FIG. 10) and containing the current injection circuitthat is powered by the leakage inductance energy.

FIG. 16 illustrates the key waveforms of the circuit depicted in FIG.15.

FIG. 17 is a schematic of the circuit labeled “Differential &Protection”, of FIG. 14.

FIG. 18 illustrates sine key waveforms of operation of the circuit ofFIG. 14.

FIG. 19 illustrates several key waveforms representing an operation ofthe circuit depicted in FIG. 13.

The sizes and relative scales of elements in Drawings may be set to bedifferent from actual size and scales to appropriately facilitatesimplicity, clarity. and understanding of the Drawings. For the samereason, not all elements present in one Drawing may necessarily be shownand/or labeled in another.

DETAILED DESCRIPTION

Utilization of various fly-back-topology-based circuits in powerconversion devices is conventionally associated with ringing, in variouswaveforms characterizing the operation of a given power-conversiondevice, caused by the energy of leakage capacitance, which energyremains substantially lost. Embodiments of the invention address thisproblem of wasting such electromagnetic energy by devising subtlemodifications of flyback-topology-based circuits that not only reducesuch circulation but also—optionally—redirect this energy to be re-usedby the conversion device.

FIG. 1 illustrates a simplified schematic of electronic circuitry of apower converter 100 configured according to a flyback topology. Suchflyback converter is formed by a transformer (Tr, 20) that has a primarywinding 22 (with corresponding inductance L1 and N1 turns in thecorresponding coil) on the primary side, and a secondary winding 24(with corresponding inductance L2 and N2 turns in the correspondingcoil); a primary or main switch (M1, 28) that is controlled by a controlvoltage signal (VcM1, 30) on the primary side. The flyback converteralso includes a parasitic capacitance (Ceq, 32) that represents thetotal parasitic capacitance reflected across the primary switch and isdisposed, on the primary side, between a terminal of the primary winding(L1, 22) and the ground. The source of the input voltage labelled as Vinor 38 is connected to another terminal of the primary winding.

The converter also includes a synchronous rectifier (SR, 34) on thesecondary side that is controlled by a control voltage signal (VcSR,25), and an output capacitor (Co, 36) disposed between the ground and aterminal of the secondary winding (L2, 24). The output voltage signalV_(o) can be read across the capacitor C_(o). In the following, any ofthe primary winding(s) and secondary winding(s) are discussed aspossessing a corresponding inductance. The terms “main switch ” and“primary switch” may be used interchangeably.

FIG. 2 illustrates the plots representing the key waveforms of a flybackconverter operating in discontinuous mode. These key waveforms includethe control signal VcM1 for the main switch M1; the current I_(M1)through the main switch; the voltage V_(M) across the main switch ; andthe voltage V_(SR) across the synchronous rectifier (SR, 34) . Duringthe operation of the circuitry 100 of FIG. 1, There exists first energycontained in the parasitic elements of the circuit (such as the leakageinductance, for example), which creates a ringing across the primaryswitch, shown as region 52 of the V_(M1) signal in FIG. 2. In aconventional flyback topology, such first energy is dissipated as is thesecond energy contained in the ringing portion 44 of the V_(M1) signalacross the main switch during the dead time period t2-t3. The ringingportion 44 defines/contains the second energy at frequencies that arelower than frequencies corresponding to the ringing 52 and the firstenergy. In addition, there exists third energy contained in theparasitic capacitance (Ceq, 32) reflected across the primary switch 28.In most of the conventional flyback converters, such energy is alsodissipated (and, therefore, lost), during the operation.

For example, in a conventional 65 W flyback converter, configuredaccording to the schematic 100 and having a leakage inductance of 1.5 uHand operating at 150 kHz, the energy contained in the leakage inductanceat full load is about 6.8 uJ.

The second energy, corresponding to the lower frequency ringing 44across the main switch after the energy is fully delivered to thesecondary side, is dependent on the input voltage 38 reflected in theprimary side. Fora parasitic capacitance Ceq of 260 pF, such secondenergy is about 4.8 uJ, and for an input voltage of 372 Vdc the secondenergy is about 11 uJ.

The energy contained in the parasitic capacitance Ceq across the primaryswitch is also function of the input voltage 38. For a parasiticcapacitance of 260 pF and DC input voltage 38 of about 141 Vdc, theenergy in Ceq is 2.58 uJ, while for a DC input voltage (V_(in),38) ofabout 373 Vdc the energy contained in Ceq is about 18 uJ. This energy isdissipated if the turning “on” of the main switch (M1, 28) is done inhard switch mode. In most of the flyback converters this energy isdissipated. For example at high input voltage of 373 Vdc and anoperation frequency of 125 KHz, the total power dissipated in theparasitic elements is about 4.43 W, which represents 6.8% of the totalprocessed power. In addition, in hard switching mode there exists aringing across the synchronized rectifier 88 (as is depicted in FIG. 2at the beginning of the time period t3-t4). The ringing across thesynchronized rectifier 88 increases the noise, and negatively impactsthe electromagnetic interference (EMI, as understood in the art) causingthe requirement for snubbers and the use of a higher voltagesynchronized rectification.

The following disclosure discusses solutions to existing problemspersisting in the process of harvesting the energy from the leakageinductance.

EXAMPLE 1 Flyback Converter with Active Clamp

One attempt for harvesting the energy of the leakage inductance,provided in related art, was to use an active clamp, as discussed in1993 in U.S. Pat. No. 5,434,768. FIG. 3 illustrates the attempt of US5,434,768 as applied to a flyback converter. Besides the general flybackcircuitry 100 presented in FIG. 1, the circuitry 300 adds a clamp (orcomplementary) switch (M2, 40), controlled by a control voltage signal(VcM2, 42), as well as a clamp capacitor (Cr, 43), together comprisingan active clamp designated as 310 and electrically connected between oneterminal of the primary winding L1 and the ground.

The mode of operation of the active-clamp-containing flyback convertercircuitry 300 is presented in FIG. 4, illustrating the key waveforms ofthe flyback circuit with the active clamp of FIG. 3. These waveformsinclude: 1) the control signal V_(C)M1 (for the main switch M1); 2) thevoltage across the main switch, VdcM1; the current I_(L1) through theprimary winding 22 of the transformer (Tr, 200; 3) the current I_(Cr)through the clamp circuit; 4) the current I_(SR) through thesynchronized rectifier ISR; and 5) the control voltage signal for theclamp (Mosfet, M2), VcM2.

Considering the time-line, between the moments t0 and t1, the mainswitch M1 is switched on, and the current I_(L1) starts to build upthrough the magnetizing inductance, thereby storing energy in thetransformer. At the moment t1, the primary switch turns off, and aresult the magnetizing current starts flowing into the secondarywinding. The leakage inductance reported to the primary side and theclamp capacitor form a resonant circuit. The current through the leakageinductance starts flowing through the clamp capacitor and the resonantcircuit, formed by the leakage inductance, and the clamp capacitorshapes the current through the clamp circuit formed by M2 and Cr,accordingly. As can be seen from the schematic of FIG. 4, the currentI_(Cr) through Cr is characterized by ringing at the frequency(ies)determined by the resonant frequency between the leakage inductance andthe clamp capacitor.

Initially, the I_(Cr) current flows through the clamp capacitor Cr,decaying and substantially reaching a zero level at the moment t2.Between the time moments t2 and t3, the current through Cr becomesnegative (which means that the current will be transferred to thesecondary side, as depicted by the curve I_(SR)-representing the currentthrough SR. Between the moments t3 and t4, the current I_(Cr) throughthe clamp circuit turns positive again while reaching the zero levelagain at about t4. After t4 the current through the clamp circuit turnsnegative and, when the clamp switch M2 turns off at t5, the energycontained in the magnetizing current adds to the existing energycontained in the resonant circuit that is formed by the primary winding(L₁, 22) and the parasitic capacitance C_(eq) reflected across the mainswitch. At the moment t5, the current through the clamp circuit is shownto have the negative amplitude (If, 46), and this current increases theamplitude of ringing during the dead time of the flyback converter (asshown in the curve V_(dc)M1) from 48 to 50. The number of ringing cyclesor undulations, as well as the polarity of the current passing throughCr, is a function of time at which M2 is switched “on” and the resonantfrequency formed by leakage inductance and the clamp capacitor Cr.Accordingly, the number of such undulations and/or polarity of thecurrent through Cr may vary. The clamp circuit (formed by M2 and Cr)takes the leakage inductance energy initially by charging the clampcapacitor Cr and further some of the energy is transferred to thesecondary side while some of the energy is bounced back (to the primaryside) and forth before the active clamp switch turns off. At the momentt5, when the clamp switch turns off, the energy remained in themagnetizing inductance adds up to the energy contained in the resonantcircuit formed by the inductance of the primary winding and theparasitic capacitance, Ceq, 32, reflected across the main switch (M1,28).

In some applications of the related art, the negative current (If, 46)passing through the clamp capacitor at the time t5, when the clampswitch turns off, is tailored to add to the energy in the resonantcircuit formed by the inductance of the primary winding of thetransformer and the parasite capacitance Ceq, in order to increase theringing amplitude and by using the valley detection approach to turn onthe main switch at a lower voltage level (or even at a zero voltagelevel) if the flyback operates in critical conduction. (A person ofskill in the art will readily appreciate that the approach that is knownas “valley detection” includes identification of the valley(s) or theportion(s) of the curve VdcM1 around the local minimum(s) of the ringingafter the moment t5 and turning the main switch “on” at the momentscorresponding to these valley(s) to reduce the main switch losses)

This way, some of the energy from the leakage inductance is used todecrease the switching losses or even eliminate them if the flybackoperates in critical conduction mode, where the main switch M1 turns onat the first “valley” (or minimum) of the ringing portion of the signal.

As was already alluded to above, the leakage inductance energy istypically lost in conventionally-configured circuitry.

EXAMPLE 2 Partial Time Active Clamp Flyback

One concept of harvesting some of the leakage inductance energy forobtaining zero voltage switching conditions for the main switch M1 hasbeen addressed U.S. patent application Ser. No. 14/933,476 titled“Partial time active clamp flyback”, the disclosure of which isincorporated herein by reference. The waveforms depicted in FIG. 5reflect the concept presented in U.S. Ser. No. 14/933,476.

The energy contained in the magnetizing current at the moment t5, whenthe active clamp switch turns off, is added to the energy of themagnetizing inductance (which is harvesting the energy of the resonantcircuit formed by the inductance of the primary winding and theparasitic capacitance Ceq). According to this concept, the energyassociated with the negative value (If, 46) of current I_(Cr) at t5 isstored and later utilized before the turn on of the main switch, tolower the voltage across the main switch M1 and to reduce theswitchinglosses.

FIG. 6 presents the plots illustrating that the “on” time for the clampcircuit is extended after the current through the synchronized rectifierreaches a zero level at t4 (see labels 48, 93). Here, the negativecurrent through the clamp capacitor Cr is increasing in amplitude duringthe time-period between t4 and t5, and reaching the value (If, 47) atthe moment t5. (Notably, the implementation of the technologicalapproach described in the patent application Ser. No. 14/933,476 canfurther add to the energy in the magnetizing inductance, and zerovoltage switching for the main switch M1 can be accomplished.)

According to the method of operation, of the flyback powerconverter-based circuitry, that is configured per the implementationdepicted in FIG. 6, the clamp control signal (VcM2, 42) has the extended“on” time (a period from t4 to t5, as shown) in order to increase theamplitude of the current (I′, 47) at the moment t5 and inject energycarried by such increased-amplitude current into the natural ringing,causing the increase of the amplitude of such ringing from thatcorresponding to the portion 48 (of curve VdsM1) to that correspondingto the portion 93 (of curve VdsM1). (As discussed further in referenceto FIGS. 10, 11, in a related embodiment of the invention, the currentthrough the clamp circuit formed by (M2, 40) and (Cr, 43), is zero atthe time when the current Icr through (SR, 34) reaches zero.)

In the event the flyback converter operates in critical mode (namely, amode in which the mains switch turns “on” during the time of the firstvalley of the VdsM1 signal, the voltage on the first valley can be zero,creating zero voltage switching condition for the main switch M1. If theflyback converter, on the other hand, operates in a conventional mode,the additional energy injection from If will increase the amplitude ofthe ringing—as depicted by the curve VdsM1 of FIG. 6, where both theinitial ringing portion 48 and the increased-in-amplitude ringingportion of the curve are shown as 48 and 93, respectively.

The drawback of the approach of U.S. Ser. No. 14/933,476 is that theleakage inductance energy circulating through the clamp is increasing,which in turn increases the conduction losses, and that the additionalenergy injection in the resonant circuit (formed by the inductance ofthe primary winding of the transformer and the parasitic capacitanceCeq) is partially dissipated due to the increased AC-impedance of thetransformer winding at the frequency of the ringing.

As follows from the above discussions of various circuits, the energyform the leakage inductance is circulating through the clamp circuit andthat circulation increases the conduction losses through the clampswitch M2. In the case wherein the “on” time of the clamp switch isextended (as depicted in FIG. 6), the energy in the clamp circuit isfurther increased and that occurrence increases the conduction losses inM2, with the benefit of increasing the energy contained in themagnetizing inductance at the end of the clamp switch M2 period ofconduction. In some application wherein the flyback operates in criticalconduction zero voltage switching can be accomplished.

EXAMPLE 3

In contradistinction with the related art, one idea of the presentinvention is implemented to avoid injecting any energy into themagnetizing inductance to avoid additional circulating current in theconverter, and stems from the realization that the minimization of thecurrent circulating through the clamp circuit formed by M2 and Cr (and,therefore, the minimization of the circulating through the clamp circuitenergy from the leakage inductance) involved reducing the currentthrough the clamp capacitor to zero at the time when the SR at thesecondary side turns “off”.

In this embodiment, the operation of which is discussed in reference toFIG. 7, the energy transferred from the clamp circuit portion (of theoverall flyback-converter-with-an-active-clamp circuitry 300 of FIG. 3)to the secondary side is caused to end before the moment of time whenthe current through the synchronized rectifier at the secondary sidereaches zero. Here, there is no additional energy injection into themagnetizing inductance after the current through the synchronizedrectification reaches zero. The energy contained in the leakageinductance is fully delivered to the secondary side and there is nofurther energy transferred to the magnetizing inductance when the clampcircuit turns off.

In this mode of operation, as is schematically depicted in FIG. 7, theclamp switch M2 is turned “off” substantially at any time between themoments t3 and t4 when the current through the clamp circuit is positive(flowing from the clamp capacitor C_(r) towards the secondary side). Asshown in FIG. 7, the clamp switch M2 is turned “off” at the moment t4.The current through the synchronized rectifier (SR, 34) continues toflow until t5, as shown. Because there is no energy transferred to themagnetizing inductance after the moment t5, the amplitude of the ringingportion 48 of the VdcM1 waveform is necessarily reduced as compared withthe previous case. In this embodiment, therefore, the energy circulatingthrough the clamp circuit formed by M2 and Cr is minimized, therebyreducing the conduction losses and increasing the efficiency of theoverall circuitry.

Overall, in this embodiment the leakage inductance energy is transferredto the secondary side and does not bounce back and forth through theclamp, and there is no energy left in the clamp at the time the clampswitch (M2, 40) turns off.

EXAMPLE 4

A related embodiment of a method for recycling the leakage inductanceenergy from the flyback converter electronic circuit of FIG. 3 is nowdescribed in reference to FIG. 8. In FIG. 8, the clamp switch M2 isturned on between the moments of time t2 to t3 9as shown—at t2).Initially, the leakage inductance energy is then transferred to the (Cr,43) through the body of the switch M2. The clamp switch M2 is turned onat t2 for a pre-determined period of time before the current I_(SR)through the SR reaches zero. The conservation of charge in the clampcapacitor Cr requires that the charge transferred to Cr be extracted andthat extraction will occur between t2 and t3. During that time interval,the current through the active clamp portion of the overall circuitrybecomes negative (changes the direction of flow) and energy istransferred to the secondary side.

The negative current Icr through the clamp capacitor increases theenergy in the magnetizing inductance and, as a result, that energy isfurther added to the energy already contained in the resonant circuitformed by the primary winding L1 and the parasitic capacitance Ceqacross the main switch. As a result, the ringing across the main switchduring the dead time period (t2 to t3) is increasing, as shown by achange of the portion of the plot representing the ringing portion, inthe curve VcM1, from 48 to 56. (Here, the portion 48 represents thevoltage across the main switch M1 without additional energy injectionform the clamp, and the portion 56 represents the voltage across themain switch with the energy injection form the clamp circuit.)

EXAMPLE 5

Yet another embodiment of the method for recycling the leakageinductance energy in the circuitry configured according to FIG. 3 isdescribed in reference to FIG. 9.

Here, the clamp switch (M2,40) is intentionally not activated during theperiod of time when the synchronized rectifier (SR, 34) is conducting.The leakage inductance energy is allowed to be fully transferred to (Cr,43) after the main switch (M1, 28) turns off at t1. After the currentI_(SR) through SR reaches zero, the dead time period (t3 . . . t4)starts, where there exists a low frequency ringing (910, see curveVdcM1) across the main switch (M1,28), caused by the resonant circuitformed by the parasitic capacitance reflected across the main switch,(Ceq, 32), and the inductance of the primary winding (L1, 22). Beforethe primary switch turns on again, the clamp switch M2 is caused to turnon at t4 for a predetermined period of time (as shown—between t4 andt5). During this time period, the energy transferred to the Cr betweent1 and t2 is extracted (removed, taken) out of Cr and furthertransferred to the secondary through SR. Some of the energy transferredto the Cr during the period (t1 . . . t2) will discharge the parasiticcapacitance Ceq. To soften the transition from the voltage level duringthe dead time to the voltage level of (Vin+nVo), where n is the ratio ofa number of turns or coils in the primary winding to that in thesecondary winding, the turn “on” of the switch M2 should be preferablycarried out at the moment corresponding to the peak of the low frequencyringing 910. Unlike in the case of utilizing the valley detectionmethodology (where the main switch is turned “on” at the valley orminimum of the ringing portion of the signal), with the use of thepresent embodiment the turn on of the clamp switch M2 is effectuated atthe peak of the ringing portion.

EXAMPLE 6 Recovery of Energy from Leakage Inductance

Another implementation of the idea of the invention is now discussed inreference to FIGS. 10, 11. Here, a problem of wasting electromagneticenergy, EM (of leakage inductance of the flyback with active clampcircuitry) contained in a ringing portion of the clamp circuitrywaveform and circulating through the clamp circuitry is addressed bycomplementing the flyback-with-active-clamp circuit with an auxiliaryelectronic system or component configured to store of EM energy (suchas, in one embodiment, an auxiliary source of voltage) and charging suchauxiliary system or component with this (otherwise wasted by relatedart) energy through an electronic valve attached to the clamp circuit.The act of so-charging the auxiliary EM storage system is carried out toreduce the rms current passing through the clamp portion of the overallcircuitry and, therefore, reduction of the ringing.

One example of the corresponding embodiment 1000 is containing theflyback converter with an active clamp (already referred to in FIG. 3)with modifications configured to implement the idea of the invention.

As shown, the primary side of the overall circuitry 1000 includes theprimary winding L1 of the transformer (Tr, 20), having Ni turns in itsprimary winding. One terminal of the primary winding L1 is directlyconnected to the source of the DC input voltage (Vin , 38) and anotherterminal of 11—shown as the node 1010—is connected with the active clampformed by the complementary switch (M2, 40) and the clamp capacitor Cr .The clamp switch is controlled by the source of a control voltage signal(VcM2, 42). The remaining portion of the primary side of the flybackconverter with the active clamp circuitry has been already discussed inreference to FIG. 3.

The secondary side of the overall circuitry 1000 includes the secondarywinding L2 of the transformer (Tr, 20) that has N2turns in its secondarywinding; the synchronous rectifier (SR, 34); and the remainingelectronic elements that have been already discussed in reference toFIG. 3.

As shown, in series with the clamp capacitor (Cr , 43) are placed tworectifiers (rectifier means, in this example depicted as diodes). Thefirst rectifier (D1, 60) has its anode connected to the clamp capacitor(C_(r), 43), at the node 1020 The second rectifier (D2, 62) has itsanode electrically connected to the input ground and its cathode—to theclamp capacitor (Cr, 43), at the node 1020. An energy storage (shownhere as the voltage source (Vb,64)) is further added between the groundand the cathode of the first rectifier. This electronic-circuitryaddition to the active-clamp portion of the circuit 1000 is designatedas 1030.

The key waveforms representing the operation of the circuit 1000 areschematically depicted in FIG. 11. The key waveforms include: a) thecontrol signal (VcM1, 30) of the main switch M1; b) the voltage (VdsM1)across the main switch M1; c) the current I_(LI) through the primarywinding L1 of the transformer (Tr, 20); d) the current I_(Cr) throughthe clamp capacitor Cr; e) the current I_(SR) through the synchronousrectifier (SR, 34); and f) the control signal VcM2 for the clamp switchM2.

Referring further to FIG. 11, in operation of the circuitry 1000, in thetime period between t0 and t1, the main switch M1 is configured toconduct and the magnetizing current builds up in the transformer Tr. Atthe moment t1, the main switch M1 turns off (shown as VcM1 reachingzero) and the magnetizing current starts flowing towards the secondarywinding (L2, 24) and through the synchronous rectifier (SR, 34). Thecurrent flowing through the leakage inductance of the primary sidestarts flowing through the clamp circuit formed by (M2, 40) and theclamp capacitor (Cr, 43), and then through the first rectifier (D1, 60)towards the auxiliary energy storage (Vb, 64). Unlike in the previousexamples of circuits discussed above—and in contradistinction with suchexamples—the current from the leakage inductance here is directedtowards the voltage source Vb. This directionality completely changesthe mode(s) of operation of the clamp circuit with which a skilledartisan may be familiar: a simple comparison of the shape of the plotI_(Cr) representing the current passing through the clamp capacitor (Cr,43) in case of the embodiment discussed in reference to FIG. 7, forexample, with that of FIG. 11 immediately illustrates the advantageousnature of operation of the embodiment 1000. The ringing portion,associated with the resonance between the leakage inductance and theclamp capacitor (see time period from t3 to t4) substantially disappearsand the time interval (between t1 to t2), during which the currentthrough the clamp capacitor reaches the zero level, substantiallyshortens. As a result, the charge injected into the clamp capacitor Cr(and substantially corresponding to the area under the Icr curve betweenthe moments t1 and t2) is significantly decreased in the case ofoperation of the circuitry 1000 as compared to that of the circuitry theoperation of which is described in FIG. 7. As a result, the electricalcharge that has to be extracted from the clamp capacitor in the timeperiod (between the moments t2 and t3) is decreased as well. Injecting(delivering) the current through D1 into a voltage source (auxiliarystorage of EM energy) V_(B) causes the transfer of energy to the voltagesource Vb and produces a dumping effect, substantially reducing theringing in the active clamp portion of the circuitry. In thisembodiment, all of the leakage inductance energy is transferred to thevoltage source (Vb, 64). The time interval between t1 and t2 is afunction of the voltage across Vb. For a larger value of Vb, the timeinterval between t1 to t2 decreases, comparatively. The voltage acrossVb is reflected across the voltage across the main switch, as depictedin the VdsM1 of FIG. 11 (an “overshoot” portion of the curve,characteristically not present in corresponding curves of either ofFIGS. 4, 5, 6, 7, 8, and 9). The reduction of the duration of the timeinterval (t1 . . . t2) leads to a considerable decrease of the RMScurrent through clamp circuit formed by (M2,40) and (Cr,43).

For example, in a conventional 65 W flyback converter having a leakageinductance of 1 micro-H and a clamp capacitor of 100 nF, the RMS currentIc, through the clamp circuit is about 0.313 A if and when the conceptof operation depicted in FIG. 7 is used. However—and in advantageouscomparison—if and when the flyback converter is equipped with theadditional storage circuitry 1030 and the principle of operationillustrated in FIG. 11 is carried out, for a Vb=10 V, the RMS currentI_(Cr) through the clamp circuit is decreased to about 0.185 A (that is,by at least 0.128 A or by at least 40%). With that, the power dissipatedin conduction in M2 (which is proportionate to the square of the RMScurrent) is reduced by about 5.98 times.

In the case of somewhat higher leakage inductance (such as, for example,4 micro-H) the advantageous impact of the usage of the circuitry portion1030 it is even stronger. Specifically, with the use of the principle ofoperation described in FIG. 7, the RMS current through the clamp circuitwould be about 0.527 A. However, if the hardware of FIG. 10 is usedoperated according to the principles of FIG. 11, then for a Vb of about10 V the RMS current through the clamp circuit is reduced to about 0.249A (that is, by about 52%)

In addition or alternatively, and further comparing the results ofoperation depicted in FIG. 7 with those depicted in FIG. 11 (for theembodiment 100 of FIG. 10), the time interval measured between themoment t1 and the moment t2 in FIG. 7 is about 1.12 micro-s. For thecircuitry 1000, on the other hand, the time interval (t1 to t2) isreduced to about 0.644 micro-s for V_(B) of 10 V.

It is appreciated, therefore, that the circuit 1030 is judiciouslyconfigured to improve the operation of the flyback converter with anactive clamp by substantially reducing the RMS current passing throughthe clamp capacitor.

The person of ordinary skill in the art will readily appreciate that oneadvantage of the use of the topology depicted as a circuit 1000 in FIG.10 (in reference to the corresponding key waveforms of FIG. 11) is thatthe energy circulating through the active clamp is significantly reducedas compared to operation of another implementation of the flybackconverter with an active clamp. Another advantageous feature is thatpart of the leakage inductance energy—otherwise substantially wasted andlost in circuits of related art—is practically stored and, therefore,can then be utilized. (For example in the last numerical example wherethe leakage inductance was 4 micro-H, an average current of 46 mA wasinjected into the Vb. If VB is configured as a source of bias voltage,then the power of about 460 mW was injected into the bias and can beused despite the fact that such amount of power may be higher than thepower consumption of the bias).

In operation, the proposed technical methodology increases theefficiency of the converter through the reduction of (by reducing) thecurrent circulating through the active clamp and by utilizing the energyof the leakage inductance. The higher the amplitude of the auxiliaryvoltage source, Vb, with which the clamp portion of the circuitry isequipped, the smaller the current circulating through the clamp portionof the overall circuit. Alternatively or in addition, because the timeinterval (t1 to t2) is reduced as compared with other related circuits,more of the energy from the leakage inductance is transferred to thesecondary side, thereby shaping the current through the synchronizedrectifier SR accordingly.

The hardware and principle of operation illustrated in reference toFIGS. 10, 11 can also be used in combination with other circuits—such asthe one the operation of which is presented in FIG. 9, for exampleComparing the key waveforms from FIGS. 9 and FIG. 12, it can be noticedthat the time interval t1 to t2 has been reduced, and even the amplitudeof the current I_(Cr)in the time interval (t4 to t5) has been reduced.These effects take place is due to the fact that if a smaller charge isstored in the clamp capacitor Cr, then the smaller charge has to betaken out during the period (t4 to t5).

It is appreciated, therefore, that embodiments of the invention providean electronic circuitry having primary and secondary sides and including(a) a flyback power converter containing an input voltage source; atransformer having primary and secondary windings, on the primary andsecondary sides, respectively; a main switch in series with the primarywinding on the primary side; and a synchronous rectifier in series withthe secondary winding on the secondary side, and (b) an active clampcircuit across the main switch, the active clamp circuit containing aclamp switch and the clamp capacitor in series with the clamp switch.Here, the clamp switch is configured to be turned on at a moment of timeafter the main switch is turned off; and the clamp switch is configuredto be turned off at a moment of time prior to the moment of time atwhich current passing through the secondary winding reaches a zerolevel. The electronic circuitry is characterized by a first value of rmscurrent through the clamp capacitor. IN one embodiment, such electroniccircuitry is further equipped with an auxiliary circuit that containstwo additional rectifiers connected in parallel with one another and inseries with an electronic component configured to store electromagneticenergy. Here, a cathode of a first of the two additional rectifiers isdirectly electrically connected with a cathode of the active clampcircuit; an anode of a second of the two additional rectifiers isdirectly electrically connected with the cathode of the first of the twoadditional rectifiers; a cathode of the second of the two additionalrectifiers is directly electrically connected with a first terminal ofthe electronic component; and a second terminal of said electroniccomponent is electrically connected with an anode of the second of thetwo additional rectifiers. Such auxiliary circuit is configured toreduce the rms current through the clamp capacitor from the first valueto a second value, the second value being at least 40% lower than thefirst value. In at least one implementation, the electronic component isconfigured as a voltage source (in this case, the first terminal is apositive terminal of said voltage source and the second terminal is anegative terminal of said voltage source). In any implementation wherethe electronic component is configured as a voltage source, such voltagesource may be configured to provide bias voltage to the flyback powerconverter. An implementation, in which the electronic component isconfigured as a voltage source configured to provide bias voltage to theflyback power converter, may additionally include a controlled switchplaced in parallel with said first of the two additional rectifiers andconfigured to be turned “on” and “off” to maintain such bias voltage ata substantially constant level. In some implementations, the primaryside additionally includes a driving winding, a first terminal of thedriving winding being directly connected to a terminal of the primarywinding, the second terminal of the driving winding being directlyconnected to a differential electronic circuitry (here, the differentialelectronic circuitry is configured to generate a control voltage signalto the clamp switch to drive the active clamp circuit). Anyimplementation in which the electronic component is configured as avoltage source that is a source of bias voltage of the flyback powerconverter, such source of bias voltage may include (i) a bias winding atthe secondary side of the transformer, the bias winding being coupledwith the primary winding and the driving winding of the primary side ofthe transformer; (ii) a bias synchronous rectifier connected to a firstterminal of the bias winding; and (iii) a bias capacitor connected to asecond terminal of the bias winding at a node that is grounded. Notably,the source of bias voltage may be configured to generate a bias voltagethat is substantially proportional to an output voltage of theelectronic circuitry measured between a terminal of the secondarywinding and the ground.

Embodiments additionally provide a method for operation of theelectronic circuitry having primary and secondary sides and comprising(i) a flyback power converter that includes an input voltage source; atransformer having primary and secondary windings, on the primary andsecondary sides, respectively; a main switch in series with the primarywinding on the primary side; and a synchronous rectifier in series withthe secondary winding on the secondary side, as well as (b) an activeclamp circuit across the main switch, the active clamp circuitcontaining a clamp switch and the clamp capacitor in series with theclamp switch, and which electronic circuitry is characterized by anelectrical charge, injected into the clamp capacitor after the mainswitch is turned off, in operation, such electrical charge having afirst charge value. The method for operation includes the steps of (a)electrically-connecting the electronic circuitry with an auxiliaryelectronic circuit in series with the clamp capacitor, where theauxiliary circuit contains two rectifiers and an auxiliary energystorage; and (b) directing a current, flowing through a leakageinductance of the primary side in operation of said circuitry, to flowthrough the clamp capacitor and then through a first of the tworectifiers towards the auxiliary energy storage to change said firstcharge value to a second charge value. Here, the second charge value issmaller than the first charge value, and a current passing through theclamp capacitor after the mains switch is turned off is a clampcapacitor current. The method may additionally include switching themain switch off, said switching off causing said directing.Alternatively or in addition, the method may include a step of injectingthe clamp capacitor current into the auxiliary energy storage. In anyimplementation, the electronic circuitry is characterized by period oftime during which the clamp capacitor current, passing through the clampcapacitor after the main switch is turned off, reaches zero, such periodof time having a first duration, while—with addition of the auxiliarycircuit—the directing includes changing the first duration to a secondduration, the second duration being shorter than the first duration. INany implementation, the electronic circuitry is characterized by theclamp capacitor current (passing through the clamp capacitor after themain switch is turned off) that has a first rms value, and—with additionof the auxiliary circuit—the directing includes changing the first rmsvalue to a second rms value, the second rms value being smaller than thefirst rms value. In any implementation, the step ofelectrically-connecting may include connecting an anode of the first ofthe two rectifiers and a cathode of the second of the two rectifiers tothe clamp capacitor and/or connecting the auxiliary energy storagebetween the ground and a cathode of the first of the two rectifiers,and/or connecting an anode of the second of the two rectifiers to theground. The auxiliary energy source may be configured as a voltagesource that is a source of bias voltage of the flyback power converter.Alternatively or in addition, the method may includeelectrically-connecting a controlled switch in parallel with the firstof the two rectifiers, the controlled switch configured to be turned“on” and “off” to maintain said bias voltage at a substantially constantlevel. In any implementation, the auxiliary energy source may beconfigured as a voltage source that is a source of bias voltage of theflyback power converter, in which case the source of bias voltageincludes (i) a bias winding at the secondary side of the transformer,the bias winding coupled with the primary winding and the drivingwinding of the primary side of the transformer; (ii) a bias synchronousrectifier connected to a first terminal of the bias winding; and (iii) abias capacitor connected to a second terminal of the bias winding at anode that is grounded. Optionally, in such a case, the source of biasvoltage is configured to generate a bias voltage that is substantiallyproportional to an output voltage of the electronic circuitry measuredbetween a terminal of the secondary winding and the ground.

EXAMPLE 7 Advantageous Utilization of Energy of Leakage Inductance forOperation of the Flyback Converter.

A person of ordinary skill in the art will readily appreciate that thetechnological implementation of the idea of the invention can be takenone step further, by addressing a problem of utilization (that is,practical use) of the energy of the leakage inductance now stored in theauxiliary EM-energy storage (shown in FIG. 10 as a source Vb). This way,not only the rms current through the active clamp portion of the overallcircuitry is reduced (with a corresponding reduction of the unwantedringing in key waveforms), but such energy may be re-used.

In one implementation for example, the energy from the leakageinductance (which is harvested/injected in the voltage source Vb) can beused to provide some or all of the bias power in the flyback convertercircuitry. One embodiment 1300 configured to implement such practicalsituation is presented in FIG. 13. Here, as compared with the embodiment1000 of FIG. 10, a bias winding (L3, 66) with N3 turns in thecorresponding coil, a synchronous rectifier (SRb, 68) and a biascapacitor (Cb1, 72) are added to form a portion 1310 of the electroniccircuitry 1300. (The synchronous rectifier SRb in one embodiment can beconfigured as a simple diode.) Based on such arrangement, in case wherethe current injected in the bias portion 1310 is larger than the currentrequired by the bias portion, the extra energy will be transferred tothe secondary side via the rectifier SRb.

In some adapter applications, such as the adapter with power delivery,for example, where the adapter communicates with the load and providesthe output voltage required by the load, the output voltage may oftenvary between 5 V and 20 V. If a conventional logic of theconverter-related electronic circuitry is followed, the bias winding L3would be tailored to provide a bias voltage that is proportionate withthe output voltage of the converter. However, because the output voltagemay vary in such a large range, the traditional arrangement bias windingsimply may not work. According to the idea of the present invention,therefore, the energy transferred through the clamp capacitor throughthe diode rectifier 60 is used to provide the bias power. The currentdemanded by the bias is proportionate with the repetition frequency ofoperation (because at each cycle a certain amount to energy is used todrive the switching devices), which makes this implementation verysuitable to the purpose as intended. At lighter loads in manyapplications the frequency decreases and as a result the bias currentdecreases and in this concept the current injection through D1 decreasesas well.

As shown in FIG. 13, a controlled switch (M3, 97), is added to thecircuitry that is controlled by a control signal (VcM3, 99) from thecontrol circuit 90.

In adapter application with power delivery wherein the output voltagewill vary between 5 V to 20 V and the bias winding method to deliver thebias power is not suitable, and the bias energy will come mostly fromthe clamp circuit via (D1, 60). In application with power delivery, thesynchronous rectifier (SRb, 68) may be replaced by a diode or notactivated. To regulate the (Vbias, 78) when the current injectionthrough D1 is too large, the switch M3 can be turned “on” as needed andin this way control the average current through D1. For example, todecrease the average current through D1, the switch (M3, 97) has to beturned on for a longer period of time than an extent of time duringwhich this switch is turned off. To increase the average current throughD1, (M3, 97), has to be off for a longer period of time than that duringwhich it is turned on. By tailoring the “on” time of (M3, 97),therefore, the average current through D1 can be regulated.

Notably, in further reference to FIG. 13, the control switch (SRB, 68)can provide substantially the same function as that described inreference to FIG. 6: the control signal (VcSRb, 70) is judiciouslyconfigured to extend the conduction after the current through thesynchronous rectifier reaches zero and (SR, 34) is turned on.

It is appreciated, therefore, that one embodiment of the inventionprovides an electronic circuitry having primary and secondary sides andcomprising (i) a flyback power converter that includes (i1) an inputvoltage source, (i2) a transformer having a primary winding on theprimary side and a secondary winding on the secondary side,respectively, (i3) a main switch in series with the primary winding onthe primary side, (i4) a synchronous rectifier in series with thesecondary winding on the secondary side; (ii) an active clamp circuit,across the main switch, that contains a clamp switch and the clampcapacitor in series with the clamp switch; (iii) a bias portion of theelectronic circuitry, including: (iii1) a bias winding on the secondaryside of the transformer, (iii2) a bias synchronous rectifier connectedto a first terminal of the bias winding, (iii3) a bias capacitorconnected to a second terminal of the bias winding at a first node thatis grounded; (iv) an auxiliary circuit containing two rectifiersconnected such that a cathode of a first of the two rectifiers isdirectly electrically connected with a cathode of the active clampcircuit at a second node, an anode of a second of the two additionalrectifiers is directly electrically connected with the cathode of thefirst of the two rectifiers at the second node, (v) a control switchconnected between the second node and the ground, and (vi) a controlelectronic circuitry configured to generate a control signal governingan operation of the control switch and electrically connected to each ofthe main switch, the bias synchronous rectifier, and the control switch.In one implementation, the operation of such electronic circuitry ischaracterized by a first value of rms current through the clampcapacitor and the auxiliary circuit is configured to reduce the rmscurrent through the clamp capacitor from the first value to a secondvalue, the second value being at least 40% lower than the first value.In any implementation the clamp switch may be configured a) to be turnedon at a moment of time after the main switch is turned off and b) to beturned off at a moment of time prior to the moment of time at whichcurrent passing through the secondary winding reaches a zero level. Inany implementation, a cathode of the second of the two rectifiers may bedirectly electrically connected with a terminal of the bias winding(which terminal is not directly connected to the bias synchronousrectifier) to provide, in operation of the electronic circuitry, a biasvoltage such that a level of the bias voltage is regulated as a resultof operation of the control switch. The control switch may be placed inparallel with the first of the two rectifiers and configured to beturned “on” and “off”, in operation of the electronic circuitry, tomaintain said level of the bias voltage substantially constant as afunction of time.

FIG. 19 illustrates empirically-acquired several key waveformsrepresenting an operation of the circuit depicted in FIG. 13. From thediscussion presented above, a person of skill in the art will readilyappreciate that, according to the idea of the invention, the (SRb, 68)is appropriately controlled to be “on” after the moment when (SR, 34) isturned “off”, to obtain zero voltage switching conditions across (M1,28) in a critical conduction mode of operation of the electronic circuit1300.

The waveforms in FIG. 19 are: voltage Vds across the main switch (M1,28), the control voltage (VcSRb, 70) governing the operation of SRb;(VcSR, 25), which is a control voltage signal for synchronous rectifier(SR, 34) on the secondary side; and the current through the bias winding(L3, 66). A skilled artisan will appreciate that the switch (SRb, 68) isturned “on” for a period of time that is longer than the “on” time ofthe (SR, 34). As a result, the current through the bias winding (L3, 66)starts to build up with a higher (steeper) slope. At the end of theperiod of conduction of the (SRb, 68), the current passing through (L3,66) has, therefore, a higher/larger amplitude to discharge (Ceg, 32)across the main switch M1 towards zero, as is depicted by the curve “VdsMain switch, M1, 28” of FIG. 19.

Notably, in this case—and in contradistinction with the operationdiscussed in reference to FIG. 6—the function of injecting more energyinto the natural ringing produced by the circuit (and resulting ineffect substantially similar to that discussed, in reference to FIG. 6,as a conversion of the ringing shown by the portion 48 to the portion93) is carried out by the element (SRb, 68) on the secondary side whilethe same function was previously performed with the switch (M2, 40) onthe primary side. It is understood, therefore, that when utilizing thisembodiment, the zero voltage switching is maintained in critical modeoperation without the use of energy circulating energy through theactive clamp. In the current embodiment, the switch (SRb, 68) thereforeis configured to perform several functions: not only as a rectifier forbias energy, but also as means/vehicle for additional energy injectionin the natural ringing across the main switch for the purpose ofobtaining zero voltage switching in the critical conduction mode.

Accordingly, embodiments of the invention provide a method for operatingthe electronic circuitry of FIG. 13, which method includes at least thestep of switching the bias synchronous rectifier on for a first periodof time that is longer than a second period of time to increase astarting amplitude of a current with which a parasitic capacitancereflected across the main switch is being discharged at an end of aconduction period of the bias synchronous rectifier. (Here, the secondperiod of time being a period of time during which the synchronousrectifier stays on.) Alternatively or in addition, the method maysatisfy at least one of the following conditions: a) formingzero-voltage switching conditions across the main switch in a criticalmode of operation of the electronic circuitry without the use of energycirculating through the active clamp; and b) having the bias synchronousrectifier configured to operate as a rectifier and as an energy-injectorin a ringing across the main switch.

EXAMPLE 8 Using the Leakage Inductance Energy to Obtain Zero-VoltageSwitching Conditions for the Mains Switch of the Circuit.

Building upon the previous Example, the harvested energy of the leakageinductance may be re-used to reduce (and even eliminate) one of thelargest losses in a flyback-type converter, specifically the switchinglosses associated with the hard discharge of (Ceq, 32) when the mainswitch (M1, 28) is turned on. The specific example of the circuit 1500of FIG. 15 achieves this goal: this circuitry is configured to harvest,in operation as illustrated schematically in FIG. 16, the leakageinductance energy and use such energy to discharge the parasiticcapacitance of the capacitor Ceq across the main switch in order tocreate zero voltage switching conditions for the main switch.

The embodiment 1500 configured such as to have the capacitor (Crc, 110)be charged with the harvested leakage inductance energy, and then, inoperation, the energy from Crc, 110), is further utilized to inject apulse of current into the transformer (Tr, 20) via the auxiliary(current injection) winding 102 with the purpose of discharging theparasitic capacitance (Ceg, 32). Here, the current injection circuit isformed with the use of a current injection winding (Linj, 102), acurrent injection switch (Minj,106) such as a Mosfet, controlled by acurrent injection control signal (VcMinj, 112), and an energy source(represented by charged capacitor Crc).

In addition to the capacitor Crc, in one implementation a smallercapacitor (Cinj, 104) is also added as shown. (In such implementation,the capacitance of the Cinj, capacitor is preferably at least 10 timeslower than the capacitance of the Crc.) In between (Cinj, 104) and (Crc,110), a diode (Dinj, 108) is placed with its anode connected to (Crc,110). The optional capacitor (Cinj, 104) is charged in forward mode viathe winding 102, during the period of conduction of the main switch (M1,28). The energy in this capacitor (Cinj, 104) is preferably smaller thanthe energy coming from the Crc. The main goal of optional use of theCinj is to shape the current through the current injection switch (Minj,106) to become negative before the moment of time when the (VcMinj, 112)turns off. In addition, the capacitor Cinj adds energy into the currentinjection circuit.

Accordingly, an embodiment of the invention provides an electroniccircuitry having primary and secondary sides and comprising: (a) aflyback power converter that includes an input voltage source; atransformer having a primary winding on the primary side and a secondarywinding on the secondary side, respectively; a main switch in serieswith the primary winding on the primary side; a parasitic capacitoracross the main switch; and a synchronous rectifier in series with thesecondary winding on the secondary side, (b) an active clamp circuitacross the main switch, the active clamp circuit containing a clampswitch and the clamp capacitor in series with the clamp switch, and (c)a current injection circuit including a current injection winding on thesecondary side of the transformer; a current injection switch connectedto a first terminal of the current injection circuit; and a source ofenergy between a second terminal of the current injection circuit andthe ground, where the current injection circuit is configured, inoperation of the electronic circuitry, to collect energy of a leakageinductance of the electronic circuitry and to inject this energy in aform of a pulse of current into the transformer via the currentinjection winding to discharge the parasitic capacitor to create a zerovoltage switching condition for the main switch. The electroniccircuitry may additionally include an auxiliary circuit that contains(i) two rectifiers connected such that a cathode of a first of the tworectifiers is directly electrically connected with a cathode of theactive clamp circuit at a second node, an anode of a second of the twoadditional rectifiers is directly electrically connected with thecathode of the first of the two rectifiers at the second node; and acontrol switch connected between the second node and the ground. Theembodiment of the electronic circuitry may further include a controlelectronic circuitry configured to generate a control signal governingan operation of the control switch and electrically connected to each ofthe main switch, the bias synchronous rectifier, and the control switch.In any implementation, the combination of the flyback converter with theactive clamp (and in absence of the auxiliary circuit) is characterizedby a first value of rms current through the clamp capacitor. In anyembodiment, the auxiliary circuit is configured to reduce the rmscurrent through the clamp capacitor from the first value to a secondvalue, such that the rms current passing through the clamp capacitor ofthe electronic circuit has the second value, the second value being atleast 40% lower than the first value. In any embodiment, the clampswitch may be configured i) to be turned on at a moment of time afterthe main switch is turned off and ii) to be turned off at a moment oftime prior to the moment of time at which current passing through thesecondary winding reaches a zero level. Optionally, the source of energyis configured as a first capacitor, and the current injection circuitadditionally includes an injection capacitor in parallel with the firstcapacitor, to shape, in operation of the electronic circuitry, a currentthrough the current injection switch to become negative before a momentof time when the current injection switch turns off. In at least oneimplementation, a capacitance of the first capacitor is at least 10times higher than a capacitance of the injection capacitor.

The key waveforms of the circuit of FIG. 15 are presented in FIG. 16.These key waveforms include: a) the control signal (VcM1, 30) for themain switch (M1,28) ; b) the voltage (VdsM1) across the main switch (M1,28); c) the magnetizing current through the transformer (Tr, 20); d) thevoltage V_(Cinj) across the capacitor Cinj; e) ID1, the current throughthe rectifier (D1,60), f) the current through the switch Minj, wheredisplayed is the current coming from Crc; and g) the control signal,VcMinj , for the current injection switch.

Considering the operation of the circuit 1500, at the moment t1 the mainswitch M1 turns off when the magnetizing current I_(M)Tr is at its peak.The current through the leakage inductance continues to flow via (M2,40) and (Cr, 43) and further through D1 to charge the capacitor (Crc,110). The voltage across Crc increases between the moments of time t1and t2 as depicted in FIG. 16, with the waveform VCrc due to the currentpassing through D1 being injected in the capacitor Crc.

Between the moments t2 and t3, the energy contained in the magnetizingcurrent of the transformer (Tr, 20) is transferred to the secondary sidevia (SR, 34) and then stored in capacitor (Co, 36). After the moment t3,the primary inductance of the transformer (Tr, 20) starts oscillatingwith the capacitor Ceq, as shown by the curve VdsM1 representing voltageacross the main switch M1.

At t4 (which moment coincides with the valley of such oscillation), theswitch Minj is turned on by the control signal (VcMinj, 112). When thecurrent injection Mosfet (Minj, 106) is turned on, the capacitor (Cinj,104) starts to discharge and the current through the switch Minjincreases. The leakage inductance of the transformer forms a resonantcircuit with the capacitor (Ceq, 32) and the current through Minj issubstantially sinusoidal. The voltage across (Cinj, 104) decays until itreaches the voltage level across Ccr at the moment t5.

After t5, the current injection is provided by the energy contained inthe capacitor Crc (in parallel with Cinj, in this example). As wasalready mentioned, it is preferred that the capacitance of Cinj muchsmaller that the capacitance of the Crc (one-tenth in value or evensmaller), in which case after the moment t5 most of the energy isdelivered from the capacitor Crc.

At the moment t6, the current injection current (that is, the currentthrough Minj) reaches zero. After t6, the current through Minj becomesnegative, because it flows into the Cinj capacitor charging it betweenthe moment t6 to t7, during the time when both the main switch M1 andthe current injection switch Minj are on.

At the moment t7, the current injection switch Minj is turned off by thecontrol signal VcMinj.

It is appreciated that the current injection circuit depicted in FIG. 15is also operational without the capacitor Cinj: the role of Cinj is notessential to the performance of this circuit, because most of the energyis delivered from the capacitor Crc, energy which comes from the leakageinductance energy of the transformer (Tr, 20).

In the circuit 1500 of FIG. 15, the amplitude of the current injectionis self-adjusting function of the voltage across the main switch whenMinj is turned on. Unlike other current injection circuits (in whichenergy usually comes only from Cinj circuit), in this current injectioncircuit the voltage across Ccr does not change significantly. Thecurrent injection amplitude is a function of the voltage across the mainswitch MI at the time when the current injection turns on. For lowvoltage across the main switch the amplitude of the current injection itis small. If the voltage across the main switch is high when the currentinjection turns on then the amplitude of the current injection high. Ifthe voltage across M1 is small (such as in the case of the lowest pointof the valley), the amplitude of the current injection is small, and ifthe turn “on” of Minj occurs at a higher input voltage across M1—theamplitude of the current injection increases.

As depicted in FIG. 16, the voltage across M1 decays as a result ofdischarge of (Ceq, 32) by the current injection through Minj reflectedinto the primary winding 22. At t6, the voltage across M1 is zero, whichcreates zero voltage switching conditions for M1.

In contradistinction with the circuitry of related art, where theleakage inductance energy is dissipated and there is additionaldissipation of energy contained in the parasitic capacitance (Ceq, 32),in this embodiment of the invention the leakage inductance energy isutilized to discharge the parasitic capacitance Ceq and create zerovoltage switching conditions for M1.

In the event the current injection current is small (because the voltageacross M1 it is low when the Minj is turned on), the energy injectedinto Crc would be higher than the energy taken out through the currentinjection. A diode (Dbias, 130) is placed between Crc and the biascircuit to allow the extra energy from the leakage inductance to be usedfor the bias utilization.

Accordingly, embodiments of the invention provide a method for operatingthe electronic circuitry of FIG. 15. Such method includes the steps of(a) turning the main switch off at a moment t1, when a magnetizingcurrent of the transformer is at a peak reached during of a period ofconduction of the main switch; (b) between the moment t1 and a momentt2, charging the source of energy, with a current from a leakageinductance of the electronic circuitry that has passed through theactive clamp and through the second of the two rectifiers, to increase avoltage across the source of energy; and (c) transferring energycontained in the magnetizing current to the secondary side via thesynchronous rectifier to store said energy in an output capacitordisposed between the ground and a terminal of the secondary winding—toharvest energy of leakage inductance of the electronic circuitry and touse said energy to discharge said parasitic capacitor to creasezero-voltage switching conditions for the main switch. The method mayadditionally include the step of: at a moment t3 (after the moment t2),having a substantially constant voltage across the main switch changedafter the moment t2,to an oscillating voltage and then turning thecurrent injection switch on. In any implementation, the method mayinclude the following step(s): after the current injection switch hasbeen turned on, discharging the source of energy to shape a currentpassing through the current injection switch to be substantiallysinusoidal, and switching off the current injection switch at a momentwhen the current passing through the current injection switch issubstantially zero. (The switching off may include switching off thecurrent injection switch after a moment when the current passing throughthe current injection switch became negative.)

EXAMPLE 9 Self-Driven Circuit

Referring again to the methodology described in reference to FIGS. 10,11, 12, a skilled artisan will readily appreciate that another advantageof the discussed methodology is provided by the proper use of thecontrol signal for the clamp switch (M2, 40). The clamp switch M2 isturned “on” for a constant time that is much shorter (for example, bymore than 50% or so) than the “on” time of such switch as used byrelated art, and, at the same time, the exact moment of turning theclamp switch M2 “off” is not particularly critical. Specifically, asshown below, the turn “off” of the switch M2 can be effectuated at anymoment after the moment t3, because there is no more current circulatingthrough the clamp switch M2.

As a result, the driving signal for the clamp switch can be derived veryeasily from the transformer using a self-driving approach. Thisimplementation is outlined in reference to FIG. 14.

Here, embodiment 1400 outlines a flyback converter with an active clampand the leakage-inductance-energy-harvesting circuit formed by (D1,60),(D2, 62) and the bias circuit formed by a bias winding 66, a rectifiermeans (SRI), 68), a bias capacitor (Cb1, 72). However, an additionalwinding 82 (which represents the self-driven clamp winding) has beenappropriately introduced on the primary side between a differential andprotection circuit 84 and the primary winding 22 such that thisadditional winding 82 is connected between the source of the clampswitch (M2, 40) the differential & protection circuit 84.

The differential & protection circuit 84 is appropriately configured toturn the clamp switch (M2, 40) “off” for a predetermined period of timethat is shorter than the time during which (SR, 34) is on and the “turnon” of the clamp switch (M2, 4) shall be initiated by the change ofpolarity in the transformer after the main switch (M1, 28) is turned“off”. The exact moment of the turn off the switch M2, on the otherhand, is not as critical, because in the event (M2, 40) is turned offafter the moment t3 and before the moment t4, there is no current flowthrough the (M2,40).

Due to the judicious configuration of the circuitry of FIG. 14, the fullenergy from the leakage inductance of transformer (Tr, 20) is deliveredthrough D1 to the capacitor (Cb1, 72). This energy is used for biasneeds of the converter, and the extra energy is transferred into thesecondary side capacitor Co.

In critical operation mode of a flyback converter, the main switch (M1,28) is turned “on” after the current through (SR, 34) reaches zero and(SR, 34) is turned off. At that time, the voltage across M1 startscollapsing (as depicted in curve VdsM1, at t5 of FIG. 4). The inductanceof the primary winding L1, starts resonating with the parasiticcapacitance reflected across the primary switch, Ceq, 32, as depicted bythe curves 50 and 48. In the critical conduction operation the mainswitch M1, 28 will turn on at the first valley of the ringing (seeportions 48, 50 of FIG. 4). Notably, in most of the current applicationsthe first valley of the ringing does not reach zero voltage. That mayhappen, however, if the current (If, 46) through the clamp capacitor(Cr, 43), depicted in FIGS. 4, 5 and 6, has enough amplitude to increasethe natural ringing as depicted in FIG. 6 (by portion 93 of the curveVdsM1).

Accordingly, embodiments of the invention provide an electroniccircuitry having primary and secondary sides and comprising: (a) aflyback power converter that includes an input voltage source; atransformer having a primary winding and a driving winding on theprimary side and secondary (with the primary and driving windings beingin series and connected to one another at a winding node); a main switchin series with the primary winding on the primary side and directlyconnected to the winding node; and a synchronous rectifier in serieswith the secondary winding on the secondary side; (b) an active clampcircuit across the main switch, the active clamp circuit containing aclamp switch and the clamp capacitor in series with the clamp switch;(c) a bias portion of the electronic circuitry, including a bias windingon the secondary side of the transformer; a bias synchronous rectifierconnected to a first terminal of the bias winding; a bias capacitorconnected to a second terminal of the bias winding at a first node thatis grounded; and (c) an auxiliary circuit containing two rectifiersconnected such that (c1) a cathode of a first of the two rectifiers isdirectly electrically connected with a cathode of the active clampcircuit at a second node, and (c2) an anode of a second of the twoadditional rectifiers is directly electrically connected with thecathode of the first of the two rectifiers at the second node. Here, acathode of the second of the two rectifiers is directly electricallyconnected with a terminal of the bias winding (which terminal is notdirectly connected to the bias synchronous rectifier) to provide, inoperation of the electronic circuitry, a bias voltage. The electroniccircuitry further contains a protection circuit connected in parallelwith the driving winding and configured to provide a control signalgoverning an operation of the clamp switch. In any implementation, theprotection circuit may include a low pass filter in series with thedriving winding. In any implementation, the flyback converter with theactive clamp (and in absence of the auxiliary circuit) are characterizedby a first value of rms current through the clamp capacitor. In anyimplementation, the auxiliary circuit is configured to reduce the rmscurrent through the clamp capacitor from the first value to a secondvalue, such that the rms current passing through the clamp capacitor ofthe electronic circuit has the second value, the second value being atleast 40% lower than the first value. In any implementation, the clampswitch may be configured i) to be turned on at a moment of time afterthe main switch is turned off and ii) to be turned off at a moment oftime prior to the moment of time at which current passing through thesecondary winding reaches a zero level.

FIG. 18 presented the empirically-acquired waveforms of the self-drivencircuit, which include a) the voltage VdsM1 across the main switch (M1,28), b) the control signal VcM1 of the main switch (M1, 28); and c) thevoltage VcM2 between the gate and source of the switch M2. As can beseen from these experimental data, the gate-to-source voltage VcM2across the switch M2 is decaying towards zero before the end of theconduction time for (SR, 34). (While the waveform for the (SR, 34) isnot presented for simplicity, this synchronous rectifier conducts—afterthe main switch M1 turns off—for the time that the voltage VdsM1 acrossM1 is at the Vin level, by analogy with the corresponding waveform ofFIG. 5.) In conclusion, the embodiment is configured to tailor themoment of the turn off the switch M2 i to be somewhere between t3 andt4.

FIG. 17 illustrates a schematic of a possible implementation of thecircuit 84, labelled “Differential & Protection” in FIG. 14. Here, thecapacitor (Cf, 116) and the resistor (Rf, 114) form a low pass filter toprotect against spikes and ringing across the driving winding (L4, 82),to prevent the switch M2 from turning “on” accidentally. The electronicelements (Cd, 118) and (Rd, 124) form a differential circuit designed toshape the voltage in the gate of the switch M2 (as depicted by thewaveform VcM2 of FIG. 18). The electronic elements (R13, 122) and (C12,126) form an additional low pass filter to prevent the turn “on” of M2by either noise or/and spikes of voltage from the winding L4.

To effectuate the operation of an embodiment of the invention, thejudicious use of a processor controlled by application-specificinstructions stored in a tangible memory element may be required. Thoseskilled in the art should readily appreciate that required algorithmicalfunctions, operations, and decisions may be implemented as computerprogram instructions, software, hardware, firmware or combinationsthereof. Those skilled in the art should also readily appreciate thatinstructions or programs defining the functions and elements of thepresent invention may be delivered to a processor in many forms,including, but not limited to, information permanently stored onnon-writable storage media (e.g. read-only memory devices within acomputer, such as ROM, or devices readable by a computer I/O attachment,such as CD-ROM or DVD disks), information alterably stored on writablestorage media (e.g. floppy disks, removable flash memory and harddrives) or information conveyed to a computer through communicationmedia, including wired or wireless computer networks. In addition, whilethe invention may be embodied in software, the functions necessary toimplement the invention may optionally or alternatively be embodied inpart or in whole using firmware and/or hardware components, such ascombinatorial logic, Application Specific Integrated Circuits (ASICs),Field-Programmable Gate Arrays (FPGAs) or other hardware or somecombination of hardware, software and/or firmware components.

Within this specification, embodiments have been described in a way thatenables a clear and concise specification to be written, but it isintended and will be appreciated that embodiments may be variouslycombined or separated without parting from the scope of the invention.In particular, it will be appreciated that each of the featuresdescribed herein is applicable to most if not all aspects of theinvention.

The disclosure of each of U.S. provisional patent application62/571,594, U.S. patent applications Ser. Nos. 15/825,647, 14/274,598and 14/933,476 is incorporated by reference herein.

The invention as recited in claims appended to this disclosure isintended to be assessed in light of the disclosure as a whole, includingfeatures disclosed in prior art to which reference is made.

For the purposes of this disclosure and the appended claims, the use ofthe terms “substantially”, “approximately”, “about” and similar terms inreference to a descriptor of a value, element, property orcharacteristic at hand is intended to emphasize that the value, element,property, or characteristic referred to, while not necessarily beingexactly as stated, would nevertheless be considered, for practicalpurposes, as stated by a person of skill in the art. These terms, asapplied to a specified characteristic or quality descriptor means“mostly”, “mainly”, “considerably”, “by and large”, “essentially”, “togreat or significant extent”, “largely but not necessarily wholly thesame” such as to reasonably denote language of approximation anddescribe the specified characteristic or descriptor so that its scopewould be understood by a person of ordinary skill in the art. In onespecific case, the terms “approximately”, “substantially”, and “about”,when used in reference to a numerical value, represent a range of plusor minus 20% with respect to the specified value, more preferably plusor minus 10%, even more preferably plus or minus 5%, most preferablyplus or minus 2% with respect to the specified value. As a non-limitingexample, two values being “substantially equal” to one another impliesthat the difference between the two values may be within the range of+/−20% of the value itself, preferably within the +/−10% range of thevalue itself, more preferably within the range of +/−5% of the valueitself, and even more preferably within the range of +/−2% or less ofthe value itself. The term substantially equivalent is used in the samefashion.

The use of these terms in describing a chosen characteristic or conceptneither implies nor provides any basis for indefiniteness and for addinga numerical limitation to the specified characteristic or descriptor. Asunderstood by a skilled artisan, the practical deviation of the exactvalue or characteristic of such value, element, or property from thatstated falls and may vary within a numerical range defined by anexperimental measurement error that is typical when using a measurementmethod accepted in the art for such purposes.

Modifications to, and variations of, the illustrated embodiments may bemade without departing from the inventive concepts disclosed herein.Furthermore, disclosed aspects, or portions of these aspects, may becombined in ways not listed above. Accordingly, the invention should notbe viewed as being limited to the disclosed embodiment(s). In addition,the terminology used herein is with the purpose of describing particularembodiments only, and is not intended to limit the scope of the presentinvention.

1. An electronic circuitry having primary and secondary sides andcomprising: a flyback power converter including: an input voltagesource; a transformer having primary and secondary windings, on theprimary and secondary sides, respectively; a leakage inductance betweenthe primary and secondary windings; a main switch in series with theprimary winding on the primary side, and a connection between the mainswitch and the primary winding which is referred to as a switching node;and wherein there is a parasitic capacitance reflected across said mainswitch; and a synchronous rectifier in series with the secondary windingand the output capacitor on the secondary side; an active clamp circuitacross the main switch, the active clamp circuit containing a clampswitch and a clamp capacitor in series with the clamp switch; a bodydiode part of the active clamp switch which conducts current while theclamp switch is turned off; wherein the current flowing out of theswitching node toward the active clamp circuit is referred to as apositive current, and the current flowing into the switch node isreferred to as a negative current; wherein a leakage inductance currentflows through the active clamp circuit after a time when the main switchturns off, wherein energy contained in the leakage inductance istransferred to the clamp capacitor; wherein an inductance of the primarywinding oscillates with the parasitic capacitance reflected across themain switch, and a voltage across the main switch rings as ringing, saidringing having valleys and hills wherein the voltage across the mainswitch has a lower amplitude at the valleys and a higher amplitude atthe hills; an auxiliary circuit which has first and second legs, whereinthe first leg contains a first diode with a cathode connected to theactive clamp circuit, and the second leg has a second diode with ananode connected to the cathode of the first diode and a cathodeconnected to a first terminal of an electronic component configured tostore electromagnetic energy; and wherein a second terminal of theelectronic component is connected to an anode of the first diode and isfurther connected to a source of said main switch; the primary andsecond windings have dots, which correspond to the polarity of therespective winding, and when a voltage is applied to the respectivewinding with a positive polarity at the dot of the respective winding,and a voltage is induced to all windings with the positive polarity atthe dot of each winding; wherein the primary winding has twoterminations, and a first of the terminations of the primary winding isconnected to the input voltage source and a second of the terminationsof the primary winding is connected to a drain of the main switch,wherein the primary winding has the dot at the first of the terminationswhich is connected to the input voltage source; a differential circuit,formed by a first resistor and a first capacitor, is connected across adrive winding, wherein the first capacitor is connected to a source ofthe clamp switch; in between a gate of the clamp switch and the sourceof the clamp switch, there is a second capacitor in parallel with asecond resistor; in between a common connection of the first capacitorand the first resistor and the gate of the clamp switch, there is acircuit formed by a third resistor in series with a circuit formed by afourth resistor in parallel with a third capacitor; during operationwhile the main switch is turned on, a voltage across the drive windinghas a voltage of negative polarity at a termination connected to thefirst resistor, and a voltage in the gate of the clamp switch becomesnegative, which turns off the clamp switch; at the time when the mainswitch is turned off, a positive voltage is induced at the non-dotconnection and for a first period of time the clamp switch is turned on.2. The electronic circuitry according to claim 1, wherein the clampswitch is turned on for a second period of time after the main switch isturned off.
 3. The electronic circuitry according to claim 1, whereinthe clamp switch is turned off prior to the main switch turning on. 4.The electronic circuitry according to claim 1, wherein the clamp switchis turned on for a third period of time, which allows the leakageinductance energy to be transferred to the clamp capacitor and allowsthe energy of the clamp capacitor to be transferred to a load.
 5. Theelectronic circuitry according to claim 1, wherein a portion of theenergy contained in the leakage inductance is transferred to theelectronic component configured to store electromagnetic energy andanother portion of the energy contained in the leakage inductance istransferred to the output.